1. Technical Field
The present invention relates to communications and, more particularly, wideband communication systems that require phase or amplitude time shifts to compensate for downstream delays experienced in one of a plurality of circuit paths.
2. Related Art
Modern RF transmitters for applications, such as cellular, personal, and satellite communications, employ digital modulation schemes such as frequency shift keying (FSK) and phase shift keying (PSK), often in combination with Code Division Multiple Access (CDMA) communication. Some of these communication schemes, for example the 270.83 kbit/s binary Gaussian FSK employed is the GSM cellular telephony standard, have constant envelopes and the transmitter signal, sRF(t). They can be represented mathematically assRF(t)=R cos(2πƒct+θ(t)),  (1)                where R denotes a constant amplitude, ƒc denotes the RF carrier frequency, and θ(t) denotes the information bearing part of the transmitted signal. FIG. 1 is a block diagram of an example transmitter appropriate for such constant-envelope modulation schemes, also referred to as a translational loop transmitter. In this transmitter architecture, the digital baseband data enters a digital processor that performs the necessary pulse shaping and modulation to produce an intermediate frequency (IF) carrier fIF signal. The resulting digital signal is converted to analog using a digital-to-analog converter (DAC) and a low pass filter (LPF) filters out undesired digital images of the IF signal. A translational loop, for example, a phase locked loop (PLL), then translates, or shifts, the IF signal to the desired RF frequency channel and a power amplifier (PA) delivers the appropriate transmit power to the antenna.        
FIG. 1 is a functional block diagram of a radio frequency transmitter architecture for constant-envelope modulation schemes. As may be seen, a digital processor is coupled to receive digital baseband data and produces a digital waveform characterized by an intermediate frequency and a phase. The output of the digital processor is produced to a digital-to-analog converter (DAC) that is capable of processing intermediate frequency digital data while avoiding unnecessary quantization noise to produce an analog outgoing signal to a low pass filter. The low pass filter removes harmonics of the analog output of the DAC and produces an outgoing low pass filtered signal to a translational loop that up-converts the analog signal from an intermediate frequency to a radio frequency. The phase information originally produced by the digital processor is maintained in the RF signal produced by the translational loop to a power amplifier for amplification and radiation from an antenna.
The RF transmitter architecture of FIG. 1 is simplistic and is intended to represent various embodiments of RF transmitters, including embodiments in which the processing described occurs for both in-phase and quadrature phase signal paths (I and Q signal paths).
FIG. 2 is a signal diagram that illustrates operation of a typical frequency modulated waveform. More specifically, the signal of FIG. 2 illustrates the transmission of a bit pattern that either is equal to 1 0 1 1 0 or 0 1 0 0 1. As may be seen, the information is represented by the carrier being modulated at one of two different frequencies which represent to two logic states “0” and “1”. This signal is a typical output of a power amplifier of the RF transmitter shown in FIG. 1 for constant-envelope modulation schemes.
FIG. 3 is a functional block diagram of an alternative translational loop RF transmitter. One typical application of the RF transmitter of FIG. 3 is GSM cellular telephony, though the concepts may readily be applied to other types of communication networks. In FIG. 3, it is assumed that a digital processor delivers the phase signal θ(t) to the transmitter for further processing and RF transmission. The transmitter is typically a digital baseband processor that performs the necessary pulse shaping, modulation, and interpolation filtering, followed by in-phase and quadrature digital-to-analog converters (DACs), low pass reconstruction filters (LPFs), and analog baseband mixers. A summing node combines the mixer outputs that are followed by low pass filtering. The remaining components of the transmitter are a phase and frequency detector (PFD), 26 MHz crystal reference (X-TAL), a charge pump (CP), a loop low pass filter (LOOP FILTER), a voltage controlled oscillator (VCO), a pair of offset mixers, as well as appropriate low pass filters. RF channel selection is achieved by employing a Fractional N frequency synthesizer. A qualitative description of the operation of the translational loop is as follows. It is easy to show that low pass filtering the sum of the mixing products of the baseband I & Q components with down-converted RF output I & Q components generates a 26 MHz sinusoid whose excess phase component equals the difference between the desired baseband phase signal and the RF output phase signal. The 26 MHz carrier is extracted by the PFD whose output is the phase error signal.
With proper PLL design, the closed loop tracking action causes the error signal to approach zero; hence, the phase of the RF output carrier at 900 MHz tracks the phase of the baseband signal, as desired.
Other types of digital communication schemes, such as the 3π/8 offset, 8-level PSK employed in the EDGE cellular telephony standard, have non-constant envelope and the transmitter signal, sRF(t), can therefore be represented in quadrature form assRF(t)=i(t)cos(2πƒct)+q(t)sin(2πƒct),  (2)
or, equivalently, in polar form assRF(t)=r(t)cos(2πƒct+θ(t)),  (3)
where both r(t) and θ(t) are information bearing components of the transmitted signal. The signal components r(t) and θ(t) are referred to as the envelope and phase of sRF(t), respectively.
FIG. 4 is a block diagram of an example RF transmitter architecture appropriate for non-constant-envelope modulation schemes. This type of transmitter is also referred to as a polar transmitter or an envelope elimination and restoration (EER) transmitter. As can be seen from FIG. 4, the digital baseband processor splits the signal into an envelope and phase component, converts the phase component to the RF via a translational loop and joins the signals at the RF via an amplitude modulated PA. This PA generates the signal defined in (3) by effectively multiplying the envelope signal with the translational loop output. The magnitude of the envelope component (magnitude component) serves to modulate a signal and power level of the power amplifier. FIG. 5 defines the envelope signal path and phase signal path of the RF polar transmitter of FIG. 4.
FIG. 6 shows a typical envelope signal, r(t), of an RF polar transmitter employed in an enhanced data rate GSM evolution (EDGE) cellular telephone system. FIG. 7 is a typical output of the translational loop section of the polar transmitter employed in EDGE cellular telephony. FIG. 8 is the PA output corresponding to modulating the translational loop output with the envelope signal. As is apparent, the resulting transmitter signal contains both amplitude and phase information.
FIG. 9 shows further details of an example implementation of the polar transmitter for the EDGE cellular telephony application. In this block diagram, it is assumed that a digital processor delivers the envelope and phase signals, r(t) and θ(t), to the transmitter for further processing and RF transmission. The envelope signal is operated upon by a digital processor that performs the necessary interpolation filtering and quantization prior to digital-to-analog conversion by a DAC. Undesired digital images of the envelope signal are filtered out by the LPF. After filtering, the envelope signal is used to directly modulate the PA output. The operation of the translational loop section of the transmitter is identical to the operation described in FIG. 3 for the constant-envelope transmitter. FIG. 10 shows an example of an ideal RF transmitter output signal power spectrum corresponding to the EDGE cellular telephony standard. The power spectrum is given in dB relative to the center of the signal, and the frequency scale is relative to the RF carrier frequency. In practice, the power spectrum emitted from an EDGE transmitter will not be ideal due to various imperfections in the RF transmitter circuitry. Thus, quality measures of the transmitter performance have been established as part of the EDGE standard and minimum requirements have been set. One quality measure that relates to the RF signal power spectrum is the so-called spectral mask, as shown in FIG. 11. This mask represents the maximum allowable levels of the power spectrum as a function of frequency offset from the RF carrier in order for a given transmitter to qualify for EDGE certification. For example, as can be seen from FIG. 11, at a frequency offset of 400 kHz (0.4 MHz), the maximum allowable emission level is −54 dB relative to the carrier (dBc). Another RF transmitter quality measure of the EDGE standard is the modulation accuracy, which relates the RF transmitter modulation performance to an ideal reference signal. Modulation accuracy is stated in root-mean-square (RMS) and peak values and is specified in percentage format. For a given transmitter to qualify for EDGE certification, the RMS modulation error must be less than 9% and the peak modulation error must be less than 30%.
One cause of performance degradation of RF polar transmitters is the so-called delay mismatch between the envelope signal and the phase signal. In the above discussion, particularly as stated in relation to equation (3), it has been assumed that the envelope and phase signals are matched in the time-domain when the envelope and translational loop output is joined at the RF frequency. However, in practical RF polar transmitters, the envelope and phase signals traverse the transmitter via different processing paths and thus may experience different signal delay before arriving at the PA. These delay variations are the result of CMOS fabrication process variations as well as temperature variations that affect the analog circuitry of the signal paths slightly differently. Such delay mismatch typically has severe impact on the spectral mask margin, while the modulation accuracy is affected to lesser degree.
FIG. 12 illustrates the concept of delay mismatch between the envelope and phase signals for an exemplary EDGE signal. Due to delay mismatch, the translational loop output is modulated by a time shifted envelope signal relative to the phase signal thereby causing an error in the transmitted RF signal. It should be noted that only delay mismatch between the envelope and phase signal paths has detrimental effect on the transmitted signal; any common delay along the paths does not affect the quality of the transmitted signal.
FIG. 13 shows an example RF signal output power spectrum corresponding to an EDGE signal with 20 nanosecond (nS) delay mismatch between the envelope and phase signal paths of the RF polar transmitter. Compared to the ideal spectrum shown in FIG. 10, the impact of delay mismatch is significant. In particular, it results in so-called spectral re-growth, which refers to the elevated power spectrum observed in the figure.
FIG. 14 shows several example RF signal output power spectra corresponding to an EDGE signal with various delay mismatches between the envelope and phase signal paths of the RF polar transmitter. The text in FIG. 14 indicates the particular values of the delay mismatch. FIG. 14 exemplifies the significant impact of delay mismatch; for example, with a delay mismatch of 80 nS or greater, the spectral mask requirement of −54 dBc at 400 kHz offset is not satisfied. In order to restore the RF transmitter performance, delay mismatch must be substantially canceled.
FIG. 15 shows an example in which an envelope signal is delayed by a slight amount, dt, to cause such delay cancellation. Equivalently, the delay cancellation could also occur along the phase signal path since only delay mismatch matters. The above discussions describe designs that are currently being considered and corresponding performance issues that relate to those designs. It is clear from the foregoing description of the related art, that what is needed, therefore, is a method and an apparatus that addresses the problem of delay mismatch to improve performance of polar transmitters that are presently being designed.